【资料名称】：网络分析仪基础

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网络分析仪基础1-1
Network Analyzer Basics Copyright
2000
Network Analyzer Basics
Slide 1
Welcome to Network Analyzer Basics.
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Network Analyzer Basics Copyright
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Network Analysis is NOT.…
Router
Bridge
Repeater
Hub
Your IEEE 802.3 X.25 ISDN
switched-packet data stream
is running at 147 MBPS with
a BER of 1.523 X 10 . . . -9
Slide 2
This module is not about computer networks! When the name "network analyzer" was
coined many years ago, there were no such things as computer networks. Back then,
networks always referred to electrical networks. Today, when we refer to the things
that network analyzers measure, we speak mostly about devices and components.
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What Types of Devices are Tested?
Passive Device type Active
Low Integration High
Antennas
Switches
Multiplexers
Mixers
Samplers
Multipliers
Diodes
Duplexers
Diplexers
Filters
Couplers
Bridges
Splitters, dividers
Combiners
Isolators
Circulators
Attenuators
Adapters
Opens, shorts, loads
Delay lines
Cables
Transmission lines
Waveguide
Resonators
Dielectrics
R, L, C's
RFICs
MMICs
T/R modules
Transceivers
Receivers
Tuners
Converters
VCAs
Amplifiers
VCOs
VTFs
Oscillators
Modulators
VCAtten’s
Transistors
Slide 3
Here are some examples of the types of devices that you can
test with network analyzers. They include both passive and
active devices (and some that have attributes of both). Many
of these devices need to be characterized for both linear and
nonlinear behavior. It is not possible to completely
characterize all of these devices with just one piece of test
equipment.
The next slide shows a model covering the wide range of
measurements necessary for complete linear and nonlinear
characterization of devices. This model requires a variety of
stimulus and response tools. It takes a large range of test
equipment to accomplish all of the measurements shown on
this chart. Some instruments are optimized for one test only
(like bit-error rate), while others, like network analyzers, are
much more general-purpose in nature. Network analyzers
can measure both linear and nonlinear behavior of devices,
although the measurement techniques are different
(frequency versus power sweeps for example). This module
focuses on swept-frequency and swept-power measurements
made with network analyzers
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Device Test Measurement Model
NF
Simple Stimulus type Complex
Response Complex
tool
Simpl
e
DC CW Swept Swept Noise 2-tone Multi- Complex Pulsed- Protocol
freq power tone
modulation RF
Det/Scope
Param. An.
NF Mtr.
Imped. An.
Power Mtr.
SNA
VNA
SA
VSA
84000
TG/SA
Ded. Testers
I-V
Absol.
Power
Gain/Flatness
LCR/Z
Harm. Dist.
LO stability
Image Rej.
Gain/Flat.
Phase/GD
Isolation
Rtn Ls/VSWR
Impedance
S-parameters
Compr'n
AM-PM
RFIC test
Full call
sequence
Pulsed S-parm.
Pulse profiling
BER
EVM
ACP
Regrowth
Constell.
Eye
Intermodulation
NF Distortion
Measurement
plane
Slide 4
Here is a key to many of the abbreviations used above:
Response
84000 series high-volume RFIC tester
Ded. Testers Dedicated (usually one-box) testers
VSA Vector signal analyzer
SA Spectrum analyzer
VNA Vector network analyzer
TG/SA Tracking generator/spectrum analyzer
SNA Scalar network analyzer
NF Mtr. Noise-figure meter
Imped. An. Impedance analyzer (LCR meter)
Power Mtr. Power meter
Det./Scope Diode detector/oscilloscope
Measurement
ACP Adjacent channel power
AM-PM AM to PM conversion
BER Bit-error rate
Compr'n Gain compression
Constell. Constellation diagram
EVM Error-vector magnitude
Eye Eye diagram
GD Group delay
Harm. Dist. Harmonic distortion
NF Noise figure
Regrowth Spectral regrowth
Rtn Ls Return loss
VSWR Voltage standing wave ratio
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Lightwave Analogy to RF Energy
RF
Incident
Reflected
Transmitted
Lightwave
DUT
Slide 5
One of the most fundamental concepts of high-frequency network analysis involves
incident, reflected and transmitted waves traveling along transmission lines. It is
helpful to think of traveling waves along a transmission line in terms of a lightwave
analogy. We can imagine incident light striking some optical component like a clear
lens. Some of the light is reflected off the surface of the lens, but most of the light
continues on through the lens. If the lens were made of some lossy material, then a
portion of the light could be absorbed within the lens. If the lens had mirrored surfaces,
then most of the light would be reflected and little or none would be transmitted
through the lens. This concept is valid for RF signals as well, except the
electromagnetic energy is in the RF range instead of the optical range, and our
components and circuits are electrical devices and networks instead of lenses and
mirrors.
Network analysis is concerned with the accurate measurement of the ratios of the
reflected signal to the incident signal, and the transmitted signal to the incident signal.
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• Verify specifications of “building blocks” for more
complex RF systems
• Ensure distortionless transmission
of communications signals
– linear: constant amplitude, linear phase / constant group
delay
– nonlinear: harmonics, intermodulation, compression, AMto-
PM conversion
• Ensure good match when absorbing
power (e.g., an antenna)
Why Do We Need to Test Components?
KPWR FM 97
Slide 6
Components are tested for a variety of reasons. Many components are used as
"building blocks" in more complicated RF systems. For example, in most transceivers
there are amplifiers to boost LO power to mixers, and filters to remove signal
harmonics. Often, R&D engineers need to measure these components to verify their
simulation models and their actual hardware prototypes. For component production, a
manufacturer must measure the performance of their products so they can provide
accurate specifications. This is essential so prospective customers will know how a
particular component will behave in their application.
When used in communications systems to pass signals, designers want to ensure the
component or circuit is not causing excessive signal distortion. This can be in the form
of linear distortion where flat magnitude and linear phase shift versus frequency is not
maintained over the bandwidth of interest, or in the form of nonlinear effects like
intermodulation distortion.
Often it is most important to measure how reflective a component is, to ensure that it
absorbs energy efficiently. Measuring antenna match is a good example.
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The Need for Both Magnitude and Phase
4. Time-domain
characterization
Mag
Time
5. Vector-error correction
Error
Measured
Actual
2. Complex impedance
needed to design
matching circuits
3. Complex values
needed for device
modeling
1. Complete
characterization of
linear networks
High-frequency transistor model
Collector
Base
Emitter
S21
S12
S11 S22
Slide 7
In many situations, magnitude-only data is sufficient for out needs. For example, we
may only care about the gain of an amplifier or the stop-band rejection of a filter.
However, as we will explore throughout this paper, measuring phase is a critical
element of network analysis.
Complete characterization of devices and networks involves measurement of phase as
well as magnitude. This is necessary for developing circuit models for simulation and
to design matching circuits based on conjugate-matching techniques. Time-domain
characterization requires magnitude and phase information to perform the inverse-
Fourier transform. Finally, for best measurement accuracy, phase data is required to
perform vector error correction.
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Agenda
􀁺 What measurements do we make?
􀃎 Transmission-line basics
􀃎 Reflection and transmission
parameters
􀃎 S-parameter definition
􀁺 Network analyzer hardware
􀃎 Signal separation devices
􀃎 Detection types
􀃎 Dynamic range
􀃎 T/R versus S-parameter test sets
􀁺 Error models and calibration
􀃎 Types of measurement error
􀃎 One- and two-port models
􀃎 Error-correction choices
􀃎 Basic uncertainty calculations
􀁺 Example measurements
􀁺 Appendix
Slide 8
In this section we will review reflection and transmission measurements. We will see
that transmission lines are needed to convey RF and microwave energy from one
point to another with minimal loss, that transmission lines have a characteristic
impedance, and that a termination at the end of a transmission line must match the
characteristic impedance of the line to prevent loss of energy due to reflections. We
will see how the Smith chart simplifies the process of converting reflection data to the
complex impedance of the termination. For transmission measurements, we will
discuss not only simple gain and loss but distortion introduced by linear devices. We
will introduce S-parameters and explain why they are used instead of h-, y-, or zparameters
at RF and microwave frequencies.
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Transmission Line Basics
Low frequencies
􀁺 wavelengths >> wire length
􀁺 current (I) travels down wires easily for efficient
power transmission
􀁺 measured voltage and current not dependent on
position along wire
High frequencies
&#1048698; wavelength ≈ or << length of transmission
medium
&#1048698; need transmission lines for efficient power
transmission
&#1048698; matching to characteristic impedance (Zo) is
very important for low reflection and maximum
power transfer
measured envelope voltage dependent on
+ I -
Slide 9
The need for efficient transfer of RF power is one of the main reasons behind the use
of transmission lines. At low frequencies where the wavelength of the signals are
much larger than the length of the circuit conductors, a simple wire is very useful for
carrying power. Current travels down the wire easily, and voltage and current are the
same no matter where we measure along the wire.
At high frequencies however, the wavelength of signals of interest are comparable to
or much smaller than the length of conductors. In this case, power transmission can
best be thought of in terms of traveling waves.
Of critical importance is that a lossless transmission line takes on a characteristic
impedance (Zo). In fact, an infinitely long transmission line appears to be a resistive
load! When the transmission line is terminated in its characteristic impedance,
maximum power is transferred to the load. When the termination is not Zo, the portion
of the signal which is not absorbed by the load is reflected back toward the source.
This creates a condition where the envelope voltage along the transmission line varies
with position. We will examine the incident and reflected waves on transmission lines
with different load conditions in following slides
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Transmission line Zo
• Zo determines relationship between voltage and current
waves
• Zo is a function of physical dimensions and ε
r
• Zo is usually a real impedance (e.g. 50 or 75 ohms)
characteristic impedance
for coaxial airlines (ohms)
10 20 30 40 50 60 70 80 90100
1.0
0.8
0.7
0.6
0.5
0.9
1.5
1.4
1.3
1.2
1.1
normalized values
50 ohm standard
attenuation is
lowest at 77 ohms
power handling capacity
peaks at 30 ohms
Microstrip
h
w
Coplanar
w1
w2
εr
Waveguide
Twisted-pair
Coaxial
b
a
h
Slide 10
RF transmission lines can be made in a variety of transmission media. Common
examples are coaxial, waveguide, twisted pair, coplanar, stripline and microstrip. RF
circuit design on printed-circuit boards (PCB) often use coplanar or microstrip
transmission lines. The fundamental parameter of a transmission line is its
characteristic impedance Zo. Zo describes the relationship between the voltage and
current traveling waves, and is a function of the various dimensions of the
transmission line and the dielectric constant ( ε
r) of the non-conducting material in the
transmission line. For most RF systems, Zo is either 50 or 75 ohms.
For low-power situations (cable TV, for example) coaxial transmission lines are
optimized for low loss, which works out to about 75 ohms (for coaxial transmission
lines with air dielectric). For RF and microwave communication and radar applications,
where high power is often encountered, coaxial transmission lines are designed to
have a characteristic impedance of 50 ohms, a compromise between maximum power
handling (occurring at 30 ohms) and minimum loss.
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Power Transfer Efficiency
RS
RL For complex impedances, maximum
power transfer occurs when ZL = ZS*
(conjugate match)
Maximum power is transferred when RL = RS
RL / RS
0
0.2
0.4
0.6
0.8
1
1.2
0 1 2 3 4 5 6 7 8 9 10
Load Power
(normalized)
Rs
RL
+jX
-jX
Slide 11
Before we begin our discussion about transmission lines, let us look at the condition
for maximum power transfer into a load, given a source impedance of Rs. The graph
above shows that the matched condition (RL = RS) results in the maximum power
dissipated in the load resistor. This condition is true whether the stimulus is a DC
voltage source or an RF sinusoid.
For maximum transfer of energy into a transmission line from a source or from a
transmission line to a load (the next stage of an amplifier, an antenna, etc.), the
impedance of the source and load should match the characteristic impedance of the
transmission line. In general, then, Zo is the target for input and output impedances of
devices and networks.
When the source impedance is not purely resistive, the maximum power transfer
occurs when the load impedance is equal to the complex conjugate of the source
impedance. This condition is met by reversing the sign of the imaginary part of the
impedance. For example, if RS = 0.6 + j0.3, then the complex conjugate RS* = 0.6 -
j0.3.
Sometimes the source impedance is adjusted to be the complex conjugate of the load
impedance. For example, when matching to an antenna, the load impedance is
determined by the characteristics of the antenna. A designer has to optimize the output
match of the RF amplifier over the frequency range of the antenna so that maximum
RF power is transmitted through the antenna
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Transmission Line Terminated with Zo
For reflection, a transmission line
terminated in Zo behaves like an infinitely
long transmission line
Zs = Zo
Zo
Vrefl = 0! (all the incident power
is absorbed in the load)
Vinc
Zo = characteristic
impedance of
transmission line
Slide 12
Let's review what happens when transmission lines are terminated in various
impedances, starting with a Zo load. Since a transmission line terminated in its
characteristic impedance results in maximum transfer of power to the load, there is no
reflected signal. This result is the same as if the transmission line was infinitely long. If
we were to look at the envelope of the RF signal versus distance along the
transmission line, it would be constant (no standing-wave pattern). This is because
there is energy flowing in one direction only.
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Transmission Line Terminated with
Short, Open
Zs = Zo
Vrefl
Vinc
For reflection, a transmission line
terminated in a short or open reflects
all power back to source
In-phase (0o) for open,
out-of-phase (180o) for short
Slide 13
Next, let's terminate our line in a short circuit. Since purely reactive elements cannot
dissipate any power, and there is nowhere else for the energy to go, a reflected wave
is launched back down the line toward the source. For Ohm's law to be satisfied (no
voltage across the short), this reflected wave must be equal in voltage magnitude to
the incident wave, and be 180o out of phase with it. This satisfies the condition that the
total voltage must equal zero at the plane of the short circuit. Our reflected and
incident voltage (and current) waves will be identical in magnitude but traveling in the
opposite direction.
Now let us leave our line open. This time, Ohm's law tells us that the open can support
no current. Therefore, our reflected current wave must be 180o out of phase with
respect to the incident wave (the voltage wave will be in phase with the incident wave).
This guarantees that current at the open will be zero. Again, our reflected and incident
current (and voltage) waves will be identical in magnitude, but traveling in the opposite
direction. For both the short and open cases, a standing-wave pattern will be set up on
the transmission line. The valleys will be at zero and the peaks at twice the incident
voltage level. The peaks and valleys of the short and open will be shifted in position
along the line with respect to each other, in order to satisfy Ohm's law as described
above.
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Transmission Line Terminated with 25 Ω
Vrefl
Standing wave pattern
does not go to zero as
with short or open
Zs = Zo
ZL = 25 Ω
Vinc
Slide 14
Finally, let's terminate our line with a 25 Ω resistor (an impedance between the full
reflection of an open or short circuit and the perfect termination of a 50 Ω load). Some
(but not all) of our incident energy will be absorbed in the load, and some will be
reflected back towards the source. We will find that our reflected voltage wave will
have an amplitude 1/3 that of the incident wave, and that the two waves will be 180o
out of phase at the load. The phase relationship between the incident and reflected
waves will change as a function of distance along the transmission line from the load.
The valleys of the standing-wave pattern will no longer be zero, and the peak will be
less than that of the short/open case.
The significance of standing waves should not go unnoticed. Ohm's law tells us the
complex relationship between the incident and reflected signals at the load. Assuming
a 50-ohm source, the voltage across a 25-ohm load resistor will be two thirds of the
voltage across a 50-ohm load. Hence, the voltage of the reflected signal is one third
the voltage of the incident signal and is 180o out of phase with it. However, as we
move away from the load toward the source, we find that the phase between the
incident and reflected signals changes! The vector sum of the two signals therefore
also changes along the line, producing the standing wave pattern. The apparent
impedance also changes along the line because the relative amplitude and phase of
the incident and reflected waves at any given point uniquely determine the measured
impedance. For example, if we made a measurement one quarter wavelength away
from the 25-ohm load, the results would indicate a 100-ohm load. The standing wave
pattern repeats every half wavelength, as does the apparent impedance.
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High-Frequency Device Characterization
Transmitted
Incident
TRANSMISSION
Gain / Loss
S-Parameters
S21, S12
Group
Delay
Transmission
Coefficient
Insertion
Phase
Reflected
Incident
REFLECTION
SWR
S-Parameters
S11, S22 Reflection
Coefficient
Impedance,
Admittance
R+jX,
G+jB
Return
Loss
Γ, ρ
Τ,τ
Incident
Reflected
R Transmitted B
A
A
R
=
B
R
=
Slide 15
Now that we fully understand the relationship of electromagnetic waves, we must also
recognize the terms used to describe them. Common network analyzer terminology
has the incident wave measured with the R (for reference) receiver. The reflected
wave is measured with the A receiver and the transmitted wave is measured with the
B receiver. With amplitude and phase information of these three waves, we can
quantify the reflection and transmission characteristics of our device under test (DUT).
Some of the common measured terms are scalar in nature (the phase part is ignored
or not measured), while others are vector (both magnitude and phase are measured).
For example, return loss is a scalar measurement of reflection, while impedance
results from a vector reflection measurement. Some, like group delay, are purely
phase-related measurements.
Ratioed reflection is often shown as A/R and ratioed transmission is often shown as
B/R, relating to the measurement receivers used in the network analyzer
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Reflection Parameters
∞ dB
No reflection
(ZL = Zo)
ρ
RL
VSWR
0 1
Full reflection
(ZL = open, short)
0 dB
1 ∞
=
ZL − ZO
ZL + ZO
Reflection
Coefficient =
Vreflected
Vincident
Γ = ρ Φ
Return loss = -20 log(ρ), ρ = Γ
Voltage Standing Wave
Ratio
VSWR = Emax
Emin
=
1 + ρ
1 - ρ
Emax
Emin
Slide 16
Let's now examine reflection measurements. The first term for reflected waves is
reflection coefficient gamma (Γ). Reflection coefficient is the ratio of the reflected
signal voltage to the incident signal voltage. It can be calculated as shown above by
knowing the impedances of the transmission line and the load. The magnitude portion
of gamma is called rho (ρ). A transmission line terminated in Zo will have all energy
transferred to the load; hence Vrefl = 0 and ρ = 0. When ZL is not equal to Zo , some
energy is reflected and ρ is greater than zero. When ZL is a short or open circuit, all
energy is reflected and ρ = 1. The range of possible values for ρ is therefore zero to
one.
Since it is often very convenient to show reflection on a logarithmic display, the second
way to convey reflection is return loss. Return loss is expressed in terms of dB, and is
a scalar quantity. The definition for return loss includes a negative sign so that the
return loss value is always a positive number (when measuring reflection on a network
analyzer with a log magnitude format, ignoring the minus sign gives the results in
terms of return loss). Return loss can be thought of as the number of dB that the
reflected signal is below the incident signal. Return loss varies between infinity for a Zo
impedance and 0 dB for an open or short circuit.
As we have already seen, two waves traveling in opposite directions on the same
transmission line cause a "standing wave". This condition can be measured in terms of
the voltage-standing-wave ratio (VSWR or SWR for short). VSWR is defined as the
maximum value of the RF envelope over the minimum value of the envelope. This
value can be computed as (1+ρ)/(1-ρ). VSWR can take
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Smith Chart Review
∞ →
Smith Chart maps
rectilinear
impedance
plane onto polar
plane
0 +R
+jX
-jX
Rectilinear impedance
plane
.
-90o
0o 180o +-
.2
.4
.6
.8
1.0
90o
0 ∞
Polar plane
Z L = Zo
Γ = 0
Constant X
Constant R
Smith chart
Γ
L Z = 0
=
±18
0 O 1
(short) ZL =
= 0 O Γ 1
(open)
Slide 17
Our network analyzer gives us complex reflection coefficient. However, we often want to know the
impedance of the DUT. The previous slide shows the relationship between reflection coefficient and
impedance, and we could manually perform the complex math to find the impedance. Although
programmable calculators and computers take the drudgery out of doing the math, a single number
does not always give us the complete picture. In addition, impedance almost certainly changes with
frequency, so even if we did all the math, we would end up with a table of numbers that may be
difficult to interpret.
A simple, graphical method solves this problem. Let's first plot reflection coefficient using a polar
display. For positive resistance, the absolute magnitude of Γ varies from zero (perfect load) to unity
(full reflection) at some angle. So we have a unit circle, which marks the boundary of the polar plane
shown on the slide. An open would plot at 1 ∠0o; a short at 1 ∠180o; a perfect load at the center,
and so on. How do we get from the polar data to impedance graphically? Since there is a one-to-one
correspondence between complex reflection coefficient and impedance, we can map one plane onto
the other. If we try to map the polar plane onto the rectilinear impedance plane, we find that we have
problems. First of all, the rectilinear plane does not have values to infinity. Second, circles of
constant reflection coefficient are concentric on the polar plane but not on the rectilinear plane,
making it difficult to make judgments regarding two different impedances. Finally, phase angles plot
as radii on the polar plane but plot as arcs on the rectilinear plane, making it difficult to pinpoint.
The proper solution was first used in the 1930's, when Phillip H. Smith mapped the impedance plane
onto the polar plane, creating the chart that bears his name (the venerable Smith chart). Since unity
at zero degrees on the polar plane represents infinite impedance, both plus and minus infinite
reactances, as well as infinite resistance can be plotted. On the Smith chart, the vertical lines on the
rectilinear plane that indicate values of constant resistance map to circles, and the horizontal lines
that indicate values of constant reactance map to arcs. Zo maps to the exact center of the chart.
In general, Smith charts are normalized to Zo; that is, the impedance values are divided by Zo. The
chart is then independent of the characteristic impedance of the system in question. Actual
impedance values are derived by multiplying the indicated value by Zo. For example, in a 50-ohm
system, a normalized value of 0.3 - j0.15 becomes 15 - j7.5 ohms; in a 75-ohm system, 22.5 - j11.25
ohms.
Fortunately, we no longer have to go through the exercise ourselves. Out network analyzer can
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Transmission Parameters
VTransmitted VIncident
Transmission Coefficient = Τ =
VTransmitted
VIncident
= τ∠φ
DUT
Gain (dB) = 20 Log
V Trans
V Inc
= 20 log τ
Insertion Loss (dB) = - 20 Log
V Trans
V Inc
= - 20 log τ
Slide 18
Transmission coefficient Τ is defined as the transmitted voltage divided by the incident
voltage. If |Vtrans| > |Vinc|, the DUT has gain, and if |Vtrans| < |Vinc|, the DUT exhibits
attenuation or insertion loss. When insertion loss is expressed in dB, a negative sign is
added in the definition so that the loss value is expressed as a positive number. The
phase portion of the transmission coefficient is called insertion phase.
There is more to transmission than simple gain or loss. In communications systems,
signals are time varying -- they occupy a given bandwidth and are made up of multiple
frequency components. It is important then to know to what extent the DUT alters the
makeup of the signal, thereby causing signal distortion. While we often think of
distortion as only the result of nonlinear networks, we will see shortly that linear
networks can also cause signal distortion.
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Linear Versus Nonlinear Behavior
Linear behavior:
&#1048698; input and output frequencies are
the same (no additional
frequencies created)
&#1048698; output frequency only undergoes
magnitude and phase change
f1 Frequency
Time